Digital television receiver converting vestigial-sideband signals to double-sideband AM signals before

ABSTRACT

A vestigial-sideband (VSB) signal is converted to a double-sideband amplitude-modulation signal having information that is subsequently detected in the double-sideband amplitude-modulation signal. A method for performing the conversion includes mixing the VSB signal with a first beat frequency different from a carrier frequency of the VSB signal, the first beat frequency being of such value as to generate a first mixing result that is translated in frequency to have a carrier at an offset frequency; mixing the VSB signal with a second beat frequency different from the carrier frequency of the VSB signal and from the first beat frequency, the second beat frequency being of such value as to generate a second mixing result that is translated in frequency to have a earner at the offset frequency; and combining the first and second mixing results to form the double-sideband amplitude-modulation signal with a carrier frequency at the offset frequency.

This is a divisional of application Ser. No. 09/440,469 filed Nov. 15,1999, now U.S. Pat. No. 6,687,313, which claims benefit of ProvisionalApplication No. 60/132,874 filed May 5, 1999, and ProvisionalApplication No. 60/138,108 filed Jun. 7, 1999.

The invention relates to radio receivers for receivingvestigial-sideband signals, which radio receivers are used in digitaltelevision sets, for example.

BACKGROUND OF THE INVENTION

Digital communications frequently employ vestigial-sideband (VSB)signals in which the passband response is reduced at carrier frequency.Excluding from consideration a pilot carrier added to the VSBsuppressed-carrier-AM digital television (DTV) signals transmitted inaccordance with the 1995 standard for digital television broadcastingestablished by the Advanced Television Standards Committee (ATSC), theradio-frequency spectrum of the VSB DTV signals exhibits 3 dB roll-offat a carrier frequency 310 khz from the lower frequency bound of thesix-megahertz-wide television channels. A problem with VSB signals withroll-off through carrier frequency is that the asymmetry of themodulation sidebands introduces jitter into carrier tracking that isdone using variants of the well-known Costas loop. In some digitalcommunications systems the transmitter employs filtering to eliminatemodulation sideband energy in the vicinity of the carrier frequency. TheATSC standard does not specifically provide for eliminating modulationsideband energy near the carrier frequency. Instead, a pilot carrier ofsubstantial strength is inserted into the VSB suppressed-carrier-AM DTVsignals to reduce the carrier jitter caused by modulation sidebandenergy near the carrier frequency.

The transient response of synchronous demodulation of VSB signals isnotoriously dependent on the roll-off of frequency response through thecarrier region in the final I-F signal being synchronously demodulated.

A type of radio receiver design that is employed in digital televisionsets employs a six-megahertz-wide final intermediate-frequency signalthat is offset from zero-frequency by no more than a few megaHertz. ThisVSB final I-F signal is digitized, converted to a complex digital finalI-F signal, and then synchrodyned to baseband using a digital complexmultiplier. The digital complex multiplier multiplies the complexdigital final I-F signal by a complex digital carrier to recoverin-phase and quadrature-phase baseband results of the synchrodynecarried out in the digital regime. The in-phase baseband results areused as symbol code input by the symbol decoder of the DTV receiver. Thequadrature-phase baseband results are lowpass filtered, and the lowpassfilter response is used to control the frequency and phase of localoscillations used in the down conversion to final I-F signal,implementing a procedure known as bandpass tracking. This type ofreceiver is more fully described in U.S. Pat. No. 5,479,449 issued 26Dec. 1996 to C. B. Patel and A. L. R. Limberg, entitled “DIGITAL VSBDETECTOR WITH BANDPASS PHASE TRACKER, AS FOR INCLUSION IN AN HDTVRECEIVER”, and assigned to Samsung Electronics Co., Ltd. U.S. Pat. No.5,479,449 describes the carrier of the final I-F signal being below anupper sideband that is synchronously detected in the digital regime torecover baseband symbol code. Such final I-F signal is the result of adownconversion in which a very-high-frequency (VHF)intermediate-frequency signal is heterodyned with local oscillations ofa VHF frequency below the VHF I-F signal frequency band. A final I-Fsignal with the carrier of above a lower sideband is the result of adownconversion in which a very-high-frequency (VHF)intermediate-frequency signal is heterodyned with local oscillations ofa VHF frequency above the VHF I-F signal frequency band. This isdescribed in U.S. Pat. No. 5,659,372 issued 19 Aug. 1997 to C. B. Pateland A. L. R. Limberg, entitled “DIGITAL TV DETECTOR RESPONDING TOFINAL-IF SIGNAL WITH VESTIGIAL SIDEBAND BELOW FULL SIDEBAND INFREQUENCY”, and assigned to Samsung Electronics Co., Ltd. U.S. Pat. No.5,659,372 describes the final I-F signal with the carrier above a lowersideband being synchrodyned to baseband in the digital regime to recoverbaseband symbol code.

SUMMARY OF THE INVENTION

A VSB signal is downconverted to a double-sideband amplitude-modulationfinal intermediate-frequency signal that is subsequently detected togenerate a baseband demodulation result. The carrier of the finalintermediate-frequency signal has a carrier offset from zero-frequency,which carrier offset exceeds the highest modulating frequency of the VSBsignal and is adjusted to a prescribed carrier offset value.

The downconversion to the DSB AM I-F signal is accomplished in certainembodiments of the invention by heterodyning the VSB signal with aheterodyning signal essentially consisting of first and second frequencycomponents. The first frequency component of the heterodyning signal islower in frequency than the carrier of the VSB signal by an amount equalto the carrier offset value prescribed for the final I-F signal. Thesecond frequency component of the heterodyning signal is higher infrequency than the carrier of the VSB signal by an amount equal to thecarrier offset value prescribed for the final I-F signal. In preferredones of these embodiments of the invention, the heterodyning signal isgenerated by a balanced modulator providing suppressed-carrieramplitude-modulation of oscillations supplied from a controlled localoscillator. The modulation of these local oscillations by the balancedmodulator is in response to a modulating signal of a frequency equal tothe carrier offset value prescribed for the final I-F signal. There isautomatic frequency and phase control (AFPC) of the local oscillationsthat the controlled local oscillator supplies. The AFPC is responsive tothe departure of the carrier of the final I-F signal from its prescribedvalue of offset from zero frequency. The DSB AM final I-F signal isdemodulated using an in-phase synchronous detector for recoveringbaseband symbol code and a quadrature-phase synchronous detector fordeveloping AFPC signal for the controlled local oscillator.

The downconversion to the DSB AM I-F signal is accomplished in otherembodiments of the invention by downconverting the VSB signalconventionally, to generate a VSB signal including a carrier frequencyoffset from zero frequency by an amount greater than the bandwidth ofthe VSB signal. The downconverted VSB signal is digitized. Then, thedigitized downconverted VSB signal is multiplied by a second harmonic ofthe carrier to generate another VSB signal, and the two digitized VSBsignals are added together to complete generation of the DSB AM signalin the digital regime.

BRIEF DESCRIPTION OF THE DRAWING

FIG. 1 is a conceptual block schematic diagram of apparatus fordemodulating in accordance with the method of the invention avestigial-sideband amplitude-modulation signal, which apparatus includesa phase-splitter to implement demodulation using a digital complexmultiplier.

FIG. 2A is a diagram of a reverse-frequency-spectrumlower-vestigial-sideband amplitude-modulation component of the finalintermediate-frequency signal that obtains when demodulating inaccordance with the invention, which reverse-frequency-spectrumcomponent is plotted as an ordinate against the same frequency abscissa.

FIG. 2B is a diagram of normal-frequency-spectrumupper-vestigial-sideband amplitude-modulation component of the finalintermediate-frequency signal that obtains when demodulating inaccordance with the invention, which normal-frequency-spectrum componentis plotted as another ordinate against the same frequency abscissa asthe frequency-spectrum component of FIG. 2A.

FIG. 2C is a diagram of the frequency spectrum of the double-sidebandamplitude-modulation final intermediate-frequency signal that obtainswhen demodulating in accordance with the invention, which complete DSBAM final I-F signal spectrum is plotted as yet another ordinate againstthe same frequency abscissa as the frequency-spectrum components ofFIGS. 2A and 2B.

FIG. 3 is a block schematic diagram of apparatus for demodulating avestigial-sideband amplitude-modulation signal, which apparatus embodiesthe invention.

FIG. 4 is a conceptual block schematic diagram of apparatus fordemodulating a vestigial-sideband amplitude-modulation signal inaccordance with the method of the invention, which apparatus usescomplex down conversion of the VSB AM signal to implement demodulationusing a digital complex multiplier, rather than using a phase-splitter.

FIG. 5 is a block schematic diagram of apparatus for demodulating avestigial-sideband amplitude-modulation signal, which apparatus embodiesthe invention and uses complex down conversion of the VSB AM signal toimplement demodulation using a digital complex multiplier.

FIG. 6 is a conceptual block schematic diagram of alternative apparatusfor demodulating in accordance with the method of the invention avestigial-sideband amplitude-modulation signal, which apparatus includesa phase-splitter to implement demodulation using a digital complexmultiplier.

FIG. 7 is a block schematic diagram of apparatus for demodulating avestigial-sideband amplitude-modulation signal, which apparatus embodiesthe invention.

FIG. 8 is a conceptual block schematic diagram of apparatus fordemodulating a vestigial-sideband amplitude-modulation signal inaccordance with the method of the invention, which apparatus usescomplex down conversion of the VSB AM signal to implement demodulationusing a digital complex multiplier, rather than using a phase-splitter.

FIG. 9 is a block schematic diagram of apparatus for demodulating avestigial-sideband amplitude-modulation signal, which apparatus embodiesthe invention and uses complex down conversion of the VSB AM signal toimplement demodulation using a digital complex multiplier.

FIG. 10 is a block schematic diagram of FIG. 7 apparatus fordemodulating a vestigial-sideband digital television (DTV) signal, asmodified by the introduction of NTSC sound trap filtering beforedigitization of the final intermediate-frequency signal.

FIG. 11 is a schematic diagram of trap filtering for co-channelinterfering NTSC analog television signal, which trap filtering issuitable for inclusion in the DTV signal receiver of FIG. 10.

FIG. 12 is a schematic diagram of a modification to the FIG. 3 circuitryin regard to processing the digitized final I-F signal from theanalog-to-digital converter, which modification is useful if roll-off ofchannel response in the carrier-frequency region is avoided in theintermediate-frequency amplifier chain.

DETAILED DESCRIPTION

FIG. 1 shows a portion of a VSB radio signal receiver following thecustomary gain-controlled VHF intermediate-frequency amplifier chain,which amplifier chain supplies VSB amplified VHF I-F signal to a mixer10 for downconversion to a final I-F signal. A voltage-controlledoscillator (VCO) 11 is designed for operation as a controlled localoscillator with automatic frequency and phase control of itsoscillations at a very high frequency f_(H). These oscillations aresupplied to a balanced amplitude-modulator 12 for modulation inaccordance with a prescribed final I-F carrier frequency f_(F). Thebalanced amplitude-modulator 12 supplies the mixer 10 a heterodyningsignal essentially consisting of a first component of frequency(f_(H)−f_(F)) and a second component of frequency (f_(H)+f_(F)). Themixer 10 multiplies the VSB amplified VHF I-F signal by the heterodyningsignal supplied by the amplitude-modulator 12. The resulting productoutput signal from the mixer 10 is lowpass filtered by a lowpass filter13 to separate a DSB AM final I-F signal from its image in the VHF band.

The DSB AM final I-F signal is supplied to a phase-splitter 14 thatconverts the real signal to a complex signal having real and imaginarycomponents supplied to a complex multiplier 15 as a complex multiplicandsignal. The complex multiplier 15 synchrodynes this complex multiplicandsignal with a final I-F carrier signal supplied to the complexmultiplier 15 as a complex multiplier signal. The resulting complexproduct supplied from the complex multiplier 15 has an in-phase (I)baseband component, which is a demodulation result descriptive of themodulating signal used in generating the transmitted VSB signalcurrently being received. The complex product also has aquadrature-phase (Q) baseband component, which is supplied to a lowpassfilter 16. The response of the lowpass filter 16 is applied to the VCO11 as an automatic frequency and phase control (AFPC) signal.

The response of the complex multiplier 15 to a component cos ω_(V)t ofthe VSB amplified VHF I-F signal will be calculated using the threewell-known trigonometric identities that follow.cos θ cos φ=0.5 cos (θ−φ)+0.5 cos (θ+φ)  (1)sin θ sin φ=0.5 cos (θ−φ)−0.5 cos (θ+φ)  (2)sin θ cos φ=0.5 sin (θ−φ)+0.5 sin (θ+φ)  (3)The local oscillations that the oscillator 11 supplies to the balancedamplitude-modulator 12 will be assumed to be of the form COS ω_(H)t, andthe modulating signal supplied to the balanced amplitude-modulator 12will be assumed to be of the form COS ω_(F)t. In accordance with theidentity (1), the response R₁₂ from the balanced amplitude-modulator 12is of the following form.R ₁₂=0.5 cos (ω_(H)−ω_(F))t+0.5 cos (ω_(H)+ω_(F))t  (4)Further in accordance with the identity (1), the response R₁₀ from themixer 10 is an ensemble of components each of the form in the followingequation (5). $\begin{matrix}\begin{matrix}{R_{10} = {{0.25{\cos( {\omega_{H} - \omega_{F} - \omega_{V}} )}t} + {0.25\quad{\cos( {\omega_{H} - \omega_{F} + \omega_{V}} )}t} +}} \\{{0.25\cos( {\omega_{H} + \omega_{F} - \omega_{V}} )t} + {0.25\quad{\cos( {\omega_{H} + \omega_{F} + \omega_{V}} )}\quad t}}\end{matrix} & (5)\end{matrix}$The lowpass filter 13 suppresses the high frequency terms in itsresponse R₁₃ to the mixer 10 response R₁₀, which response R₁₃ is anensemble of components each of the form in the following equation (6).R ₁₃=0.25 cos (ω_(H)−ω_(F)−ω_(V))t+0.25 cos (ω_(H)+ω_(F)−ω_(V))t  (6)The phase-splitter 14 repeats the lowpass filter 13 response R₁₃ as itsreal response Re₁₄, which is an ensemble of components each of the formin the following equation (7), and generates its imaginary responseIm₁₄, which is an ensemble of components each of the form in thefollowing equation (8).Re ₁₄=0.25 cos (ω_(H)−ω_(F)−ω_(V))t+0.25 cos (ω_(H)+ω_(F)−ω_(V))t  (7)Im ₁₄=0.25 sin (ω_(H)−ω_(F)−ω_(V))t+0.25 sin (ω_(H)+ω_(F)−ω_(V))t  (8)The in-phase response I of the complex multiplier 15 is an ensemble offrequency components, each defined by the following equations (9).$\begin{matrix}\begin{matrix}{I = {{{Re}_{14}\cos\quad\omega_{F}t} + {{Im}_{14}\sin\quad\omega_{F}t}}} \\{= {{0.25{\cos( {\omega_{H} - \omega_{F} - \omega_{V}} )}t*\cos\quad\omega_{F}t} +}} \\{{0.25{\cos( {\omega_{H} + \omega_{F} - \omega_{V}} )}t*\cos\quad\omega_{F}t} +} \\{{0.25{\sin( {\omega_{H} - \omega_{F} - \omega_{V}} )}t*\sin\quad\omega_{F}t} +} \\{0.25{\sin( {\omega_{H} + \omega_{F} - \omega_{V}} )}t*\sin\quad\omega_{F}t} \\{= {{\cos( {\omega_{H} - \omega_{V}} )}t}}\end{matrix} & (9)\end{matrix}$The quadrature-phase response Q of the complex multiplier 15 is anensemble of frequency components, each defined by the followingequations (10). $\begin{matrix}\begin{matrix}{Q = {{{Re}_{14}\sin\quad\omega_{F}t} - {{Im}_{14}\cos\quad\omega_{F}t}}} \\{= {{0.25\cos\quad( {\omega_{H} - \omega_{F} - \omega_{V}} )\quad t*\sin\quad\omega_{F}t} +}} \\{{0.25\cos\quad( {\omega_{H} + \omega_{F} - \omega_{V}} )\quad t*\sin\quad\omega_{F}t} -} \\{{0.25\sin\quad( {\omega_{H} - \omega_{F} - \omega_{V}} )\quad t*\cos\quad\omega_{F}t} -} \\{0.25\sin\quad( {\omega_{H} + \omega_{F} - \omega_{V}} )\quad t*\cos\quad\omega_{F}t} \\{= {\sin\quad( {\omega_{H} - \omega_{V}} )\quad t}}\end{matrix} & (10)\end{matrix}$The I and Q responses are the same as for the well-known Costas loopdescribed in U.S. Pat. No. 3,101,448 issued 20 Aug. 1963 to J. P. Costasand titled “SYNCHRONOUS DETECTOR SYSTEM”.

So, the design of the filter 16 in the AFPC loop controlling the VCO 11can follow convention practice for Costas loops. As one skilled in theart and acquainted with the foregoing disclosure will appreciate, thereare other circuit arrangements that use two separate local oscillatorsfor generating the two tracking frequency terms 0.5 COS (ω_(H)−ω_(F))tand 0.5 cos (ω_(H)+ω_(F))t, rather than the single local oscillator 11and the balanced amplitude-modulator 12. However, tracking theoscillators with separate AFPC loops is considerably more difficult toimplement successfully in practice.

The matter that is considered further with reference to FIGS. 2A, 2B and2C is how the VSB amplified VHF I-F signal supplied to the mixer 10 isconverted to a DSB AM signal in the response R₁₃ of the lowpass filter13 following the mixer 10. The VSB amplified VHF I-F signal usuallyexhibits a reverse-frequency-spectrum so the principal sideband is lowerin frequency than the remnant, suppressed sideband close to the VHF I-Fcarrier. This reverse-frequency-spectrum VHF I-F signal obtains if thereis a direct downconversion of the radio-frequency (R-F) VSB DTV signalby superheterodyne with a tuned oscillator of higher frequency, similarto what is done in commercial analog TV receivers. Thisreverse-frequency-spectrum VHF I-F signal also obtains in certainplural-conversion receivers. In a plural-conversion receiver of thistype the R-F VSB DTV signal is first upconverted to anultra-high-frequency intermediate frequency signal by superheterodynewith oscillations of higher frequency than its own that are suppliedfrom a tuned oscillator. In this type of plural-conversion receiver, inorder to downconvert the UHF I-F signal to the VHF I-F signal, the UHFI-F signal is subsequently heterodyned with oscillations of frequencylower than its own, which oscillations are supplied from afixed-frequency oscillator.

FIG. 2A diagrams the reverse-frequency-spectrum, principallylower-sideband component of the final I-F signal that results from thereverse-spectrum VHF I-F signal heterodyning with the 0.5 cos(ω_(H)−ω_(F))t component of the balanced amplitude-modulator 12 outputsignal R₁₂, which is lower in frequency than the VHF I-F signal. In thistype of receiver the FIG. 2A reverse-frequency spectrum is the ensembleof 0.25 cos (ω_(H)−ω_(F)−ω_(V))t terms in the lowpass filter 13 responseR₁₃ for all ω_(V) terms in the reverse-spectrum VHF I-F signal.

FIG. 2B diagrams the normal-frequency-spectrum, principallyupper-sideband amplitude-modulation component of the final I-F signalthat results from the reverse-spectrum VHF I-F signal heterodyning withthe 0.5 cos (ω_(H)−ω_(F))t component of the balanced amplitude-modulator12 output signal R₁₂, which is higher in frequency than the VHF I-Fsignal. In this type of receiver the FIG. 2B normal-frequency spectrumis the ensemble of 0.25 cos (ω_(H)+ω_(F)−ω_(V))t terms in the lowpassfilter 13 response R₁₃ for all ω_(V) terms in the reverse-spectrum VHFI-F signal.

FIG. 2C diagrams the frequency spectrum of the DSB AM final I-F signalthat the lowpass filter 13 supplies as its response R₁₃. This spectrumis the sum of the subspectra that FIGS. 2A and 2B respespectivelydiagram. The angular frequency ω_(F) in radians/second is 2π times thefinal I-F signal carrier frequency f_(F) in cycles per second that isshown on the abscissa axis for FIGS. 2A, 2B and 2C.

A normal-frequency-spectrum VHF I-F signal obtains in another type ofplural-conversion receiver. In a plural-conversion receiver of thistype, also, the R-F VSB DTV signal is first upconverted to anultra-high-frequency intermediate frequency signal by superheterodynewith oscillations of higher frequency than its own that are suppliedfrom a tuned oscillator. In this type of plural-conversion receiver,however, in order to downconvert the UHF I-F signal to the VHF I-Fsignal, the UHF I-F signal is subsequently heterodyned with oscillationsof frequency higher than its own, which oscillations are supplied from afixed-frequency oscillator. The normal-frequency-spectrum VHF I-F signalwill heterodyne with the 0.5 cos (ω_(H)+ω_(F))t component of thebalanced amplitude-modulator 12 output signal R₁₂, which is higher infrequency than the VHF I-F signal, to generate thereverse-frequency-spectrum, principally lower-sideband component of thefinal I-F signal shown in FIG. 2A. In this type of receiver the FIG. 2Areverse-frequency spectrum is the ensemble of 0.25 cos(ω_(H)−ω_(F)−ω_(V))t terms in the lowpass filter 13 response R₁₃ for allω_(V) terms in the normal-spectrum VHF I-F signal. Thenormal-frequency-spectrum VHF I-F signal will heterodyne with the 0.5cos (ω_(H)−ω_(F))t component of the balanced amplitude-modulator 12output signal R₁₂, which is lower in frequency than the VHF I-F signal,to generate the normal-frequency-spectrum, principally upper-sidebandcomponent of the final I-F signal shown in FIG. 2B. In this type ofreceiver the FIG. 2B reverse-frequency spectrum is the ensemble of 0.25cos (ω_(H)+ω_(V))t terms in the lowpass filter 13 response R₁₃ for allω_(V) terms in the normal-spectrum VHF I-F signal.

FIG. 3 shows in more detail how the FIG. 1 concept is implemented whenthe demodulation is carried out in the digital regime, with the phasesplitter 14 being a phase-splitter 014 of digital-filter type and thecomplex multiplier 15 being a digital complex multiplier 015. Thelowpass filter 13 response R₁₃ is digitized by an analog-to-digitalconverter 17, and the resulting digitized DSB AM signal is applied asinput signal to the phase-splitter 014. The in-phase response I suppliedby the digital complex multiplier 015 as the real part of the complexproduct signal therefrom is a digital baseband signal, which is suitedfor application to subsequent portions of the receiver not shown in FIG.3. These subsequent portions include baseband equalization and ghostcancellation filtering and subsequent symbol decoder apparatus. Thequadrature-phase response Q supplied by the digital complex multiplier015 as the imaginary part of the complex product signal therefrom is adigital baseband signal, which a digital-to-analog converter 19 convertsto analog form to provide the input signal to the analog lowpass filter16 that supplies AFPC signal to the VCO 11.

In the FIG. 3 apparatus variously phased digital carrier waves of ω_(F)radian/second frequency are generated from read-only memories 20, 21 and22. Sampling in the FIG. 3 apparatus is synchronized to a rationalmultiple of symbol rate, which is most expeditiously implemented asfollows, using elements not shown in FIG. 3. An envelope detector forthe amplified VSB I-F signal is used to develop an envelope detectorresponse that contains frequency components subharmonic to the symbolrate of the VSB DTV signal. The component at half symbol rate isselected by a narrowband bandpass filter and is rectified or doubled torecover symbol frequency against which a sample clock oscillator issynchronized by an automatic frequency and phase control loop. A samplecounter counts average-axis-crossings of the oscillations from thesample clock oscillator and addressing for the ROMs 20, 21 and 22 isderived from the sample count or some portion thereof.

The ROM 20 stores a look-up table of sin ω_(F)t values. The digitalsamples descriptive of the sin ω_(F)t system function are supplied to adigital-to-analog converter 23 that responds with analog sin ω_(F)tsignal supplied to a phase detector 24. The phase detector 24 comparesthis analog sin ω_(F)t signal with oscillations from avoltage-controlled oscillator 25 to generate an automatic frequency andphase control signal for the VCO 25. This AFPC signal locks the VCO 25oscillations in quadrature phase with the analog sin ω_(F)t input signalsupplied to the phase detector 24. Consequently, the VCO 25 supplies cosω_(F)t oscillations, which are applied to the balancedamplitude-modulator 12 as modulating signal.

With reasonable care in the design of the VCO 25 there is very littleharmonic distortion accompanying these cos ω_(F)t oscillations.Alternatively, a cos ω_(F)t system function could be drawn from ROM andconverted to an analog cos ω_(F)t signal to be supplied to the modulator12 as modulating signal. However, in DTV the system sampling rate is notmany times higher than the carrier frequency f_(F) of the final I-Fsignal so quantizing distortion is a problem. Analog filtering tosuppress the quantizing distortion tends to be expensive and tointroduce delay differences in the analog cos ω_(F)t signal betweenvarious receivers which complicates the mass-manufacturing of receiverswith as few production line adjustments as possible.

The ROMs 21 and 22 supply the digital complex multiplier 015 samplestreams respectively descriptive of a cos ω_(F)t system function anddescriptive of a sin ω_(F)t system function. These cos ω_(F)t and sinω_(F)t system functions are delayed to compensate for the latent delaysin the analog lowpass filter 13, the ADC 17, the phase-splitter 14, etc.

FIG. 4 shows a modification of the FIG. 1 portion of a VSB radio signalreceiver that uses a complex mixer instead of the mixer 10 fordownconverting VSB VHF I-F signal to DSB AM final I-F signal. Thisavoids the need for the phase-splitter 14 before the complex multiplier15 used for demodulation. The complex mixer comprises component mixers100 and 101 having their respective output signals filtered by lowpassfilters 130 and 131, respectively. The lowpass filters 130 and 131supply their responses to the complex multiplier 15 as real andimaginary signals, respectively. The mixers 100 and 101 receive similarVSB amplified VHF I-F signals as respective multiplicand input signalsto be downconverted to a DSB AM final I-F signal, which VSB signals canbe supplied from the customary gain-controlled VHF I-F amplifier chain.The FIG. 1 VCO 11 supplying cos ω_(H)t real or in-phase localoscillations is replaced in FIG. 4 by a VCO 011 supplying sin ω_(H)timaginary or quadrature-phase local oscillations, as well as supplyingcos ω_(H)t real or in-phase local oscillations. The COS ω_(H)t in-phaselocal oscillations from the VCO 011 are supplied to a balancedamplitude-modulator 120 there to be modulated by cos ω_(F)t modulatingsignal to generate a multiplier input signal for the component mixer100. The operation of the modulator 120, the mixer 100 and the lowpassfilter 130 in the portion of a VSB signal receiver shown in FIG. 4corresponds with the operation of the modulator 12, the mixer 10 and thelowpass filter 13 in the portion of a VSB signal receiver shown in FIG.1. Accordingly, the response R₁₃₀ is an ensemble of terms each of thefollowing form.R ₁₃₀=0.25 cos (ω_(H)−ω_(F)−ω_(V))t+0.25 cos (ω_(H)+ω_(F)−ω_(V))t  (11)The sine ω_(H)t in-phase local oscillations from the VCO 011 aresupplied to a balanced amplitude-modulator 121 there to be modulated bycos ω_(F)t modulating signal to generate a multiplier input signal forthe component mixer 101. In accordance with the trigonometric identity(3) set forth above, the response R₁₂₁ of the modulator 121 is of thefollowing form.R ₁₂₁=0.5 sin (ω_(H)−ω_(F))t+0.5 sin (ω_(H)+ω_(F))t  (12)Further in accordance with the identity (3), the product output responseR₁₀₁ from the mixer 101 to the multiplication of a cos ω_(H)tmultiplicand input signal by the R₁₂₁ multiplier input signal is anensemble of terms each of the following form. $\begin{matrix}\begin{matrix}{R_{101} = {{0.25{\sin( {\omega_{H} - \omega_{F} - \omega_{V}} )}\quad t} + {0.25\quad{\sin( {\omega_{H} - \omega_{F} + \omega_{V}} )}\quad t} +}} \\{{0.25{\sin( {\omega_{H} + \omega_{F} - \omega_{V}} )}\quad t} + {0.25\quad{\sin( {\omega_{H} + \omega_{F} + \omega_{V}} )}\quad t}}\end{matrix} & (13)\end{matrix}$The lowpass filter 131 suppresses the high frequency terms in itsresponse R₁₃₁ to the mixer 101 response R₁₀₁. Accordingly, the responseR₁₃₁ is an ensemble of terms each of the following form. R ₁₃₁=0.25 sin (ω_(H)−ω_(F)−ω_(V))t+0.25 sin (ω_(H)+ω_(F)−ω_(V))t  (14)The lowpass filter 131 response R₁₃₁ in equation (14) preceding is thesame as the imaginary response Im₁₄ of the phase-splitter 14 as setforth in equation (8) above, it is noted.

FIG. 5 shows modifications of the FIG. 3 portion of the VSB signalreceiver to use the complex mixer composed of component mixers 100 and101 rather than the mixer 10, so that the phase-splitter 014 ofdigital-filter type is not required for supplying complex multiplicandinput signal to the digital complex multiplier 015. As in FIG. 4, FIG. 5shows the component mixers 100 and 101 having their respective outputsignals filtered by lowpass filters 130 and 131, respectively. Themixers 100 and 101 receive similar VSB amplified VHF I-F signals asrespective multiplicand input signals to be downconverted to a DSB AMfinal I-F signal, which VSB signals can be supplied from the customarygain-controlled VHF I-F amplifier chain. The FIG. 3 VCO 11 supplying cosω_(H)t real or in-phase local oscillations is replaced in FIG. 5 by theVCO 011 supplying sin ω_(H)t imaginary or quadrature-phase localoscillations, as well as supplying cos ω_(H)t real or in-phase localoscillations. The cos ω_(H)t in-phase local oscillations from the VCO011 are supplied to the balanced amplitude-modulator 120 there to bemodulated by cos ω_(F)t modulating signal to generate a multiplier inputsignal for the component mixer 100. The operation of the modulator 120,the mixer 100 and the lowpass filter 130 in the portion of a VSB signalreceiver shown in FIG. 5 corresponds with the operation of the modulator12, the mixer 10 and the lowpass filter 13 in the portion of a VSBsignal receiver shown in FIG. 3. As in FIG. 4, FIG. 5 shows the sineω_(H)t in-phase local oscillations from the VCO 011 being supplied tothe balanced amplitude-modulator 121 there to be modulated by cos ω_(F)tmodulating signal to generate the multiplier input signal for thecomponent mixer 101.

The responses R₁₃₀ and R₁₃₁ of the lowpass filters 130 and 131 aredigitized by analog-to-digital converters 170 and 171, respectively, andthe resulting real and imaginary components of the digitized DSB AMsignal are applied to the digital complex multiplier 015 as real andimaginary signals, respectively. The in-phase response I supplied by thedigital complex multiplier 015 as the real part of the complex productsignal therefrom is a digital baseband signal, which is suited forapplication to subsequent portions of the receiver not shown in FIG. 5.These subsequent portions include baseband equalization and ghostcancellation filtering and subsequent symbol decoder apparatus. Thequadrature-phase response Q supplied by the digital complex multiplier015 as the imaginary part of the complex product signal therefrom is adigital baseband signal, which a digital-to-analog converter 19 convertsto analog form to provide the input signal to the analog lowpass filter16 that supplies AFPC signal to the VCO 011.

The ROMs 20, 21 and 22 in the FIG. 5 apparatus are connected andoperated the same as in the FIG. 3 apparatus. The connection and theoperation of elements 20, 23, 24 and 25 to generate a cos ω_(F)tmodulating signal for the balanced amplitude-modulator 12 is the same inthe FIG. 5 apparatus as in the FIG. 3 apparatus.

The invention as thusfar described converts the vestigial sideband (VSB)signal to a double-sideband amplitude-modulation signal in the analogregime, then digitizes the resulting DSB AM signal and demodulates thedigitized DSB AM signal in the digital regime. Such arrangements requirethe analog mixer 10 (or each of the analog mixers 100 and 101 in acomplex downconversion) to have good linearity with regard both tomultiplier and multiplicand input signals. Switching converters are notpossible if the multiplier signal in the downconversion comprises morethan one carrier frequency.

In the embodiments of the invention described following, the DSB AMsignal is generated in the digital regime, proceeding from the VSBsignal as downconverted to include a carrier frequency offset from zerofrequency by an amount greater than the bandwidth of the VSB signal. Thedownconverted VSB signal is digitized. Then, the digitized downconvertedVSB signal is multiplied by a second harmonic of the carrier to generateanother VSB signal, and the two digitized VSB signals are added togetherto complete generation of the DSB AM signal in the digital regime. Thereare embodiments of the invention in which the downconversion in theanalog regime of the VSB signal to final I-F signal is done so as torecover the reversed frequency spectrum lower sideband shown in FIG. 2A,with the normal frequency spectrum upper sideband shown in FIG. 2B beingcreated during digital processing. However, in the preferred embodimentsof the invention described hereinafter, the downconversion in the analogregime of the VSB signal to final I-F signal is done so as to recoverthe normal frequency spectrum upper sideband shown in FIG. 2B, with thereversed frequency spectrum lower sideband shown in FIG. 2A beingcreated during digital processing. This latter scheme for performingdownconversion in the analog regime facilitates filtering to trap NTSCaudio signals that might otherwise interfere with DTV reception.

FIG. 6 shows a portion of a VSB radio signal receiver following thecustomary gain-controlled VHF intermediate-frequency amplifier chain,which differs from the portion of a VSB radio signal receiver shown inFIG. 1 in the following ways. The VCO 11 supplies its oscillations at avery high frequency f_(H) directly to the mixer 10 without theinterposition of the balanced amplitude-modulator 12 of FIG. 1. Themixer 10 multiplies the VSB amplified VHF I-F signal by the heterodyningsignal of frequency f_(H) and the resulting product output signal fromthe mixer 10 is lowpass filtered by a lowpass filter 13 to separate aVSB final I-F signal with carrier frequency f_(f) from its image in theVHF band. An adder 26 generates DSB AM signal as its sum output signalby combining two VSB signals received as summand input signals, one VSBsignal providing the lower sideband of that DSB AM signal, and the otherVSB signal providing the upper sideband of that DSB AM signal. Thelowpass filter 13 response is one of the two summand input signals ofthe adder 26. The other summand input signal of the adder 26 is theproduct output signal of a balanced amplitude modulator 27 whichmodulates a suppressed carrier frequency 2f_(f) in accordance with thelowpass filter 13 response. The DSB AM signal that the adder 26generates as its sum output signal is supplied to the phase-splitter 14to generate the complex samples of the DSB AM signal that the complexmultiplier 15 demodulates to recover baseband symbol code as an in-phasedemodulation result and to recover AFPC loop error signal as aquadrature-phase demodulation result.

In the FIG. 6 downconversion circuitry, in accordance with thetrigonometric identity (1), the response R₁₀ from the mixer 10 will bean ensemble of terms each of the following form, presuming the VCO 11 tobe of the form COS ω_(H)t.R ₁₀=0.5 cos (ω_(H)−ω_(V))t+0.5 cos (ω_(H)+ω_(V))t  (15)The lowpass filter 13 suppresses the high frequency terms in itsresponse R₁₃ to the mixer 10 response R₁₀, to generate an ensemble ofterms each per the following equation (16).R ₁₃=0.5 cos (ω_(H)−ω_(V))t  (16)The balanced amplitude modulator 27 modulates a suppressed 2 cos 2ω_(F)t carrier by the lowpass filter 13 response R₁₃ to generate in itsresponse R₂₇, in accordance with the trigonometric identity (1), anensemble of terms each per the following equation (17).R ₂₇=0.5 cos (2ω_(F)+ω_(H)−ω_(V))t+0.5 cos (2ω_(F)−ω_(H)+ω_(V))t  (17)The adder 26 sums R₁₃ and R₂₇ to generate a sum output signal R₂₆ whichis an ensemble of terms each per the following equation (18).$\begin{matrix}\begin{matrix}{R_{26} = {R_{13} + R_{27}}} \\{{= {{0.5{\cos( {\omega_{H} - \omega_{V}} )}\quad t} +}}\quad} \\{{{0.5{\cos( {{2\omega_{F}} + \omega_{H} - \omega_{V}} )}\quad t} + {0.5{\cos( {{2\omega_{F}} - \omega_{H} + \omega_{V}} )}t}}\quad}\end{matrix} & (18)\end{matrix}$The phase-splitter 14 repeats the adder 26 response R₂₆ as its realresponse Re₁₄, an ensemble of terms each per the following equation(19), and generates its imaginary response Im₁₄, an ensemble ofcorresponding terms each per the following equation (20).$\begin{matrix}\begin{matrix}{{{Re}_{14} = {{0.5{\cos( {\omega_{H} - \omega_{V}} )}\quad t} +}}\quad} \\{{0.5{\cos( {{2\omega_{F}} + \omega_{H} - \omega_{V}} )}\quad t} + {0.5{\cos( {{2\omega_{F}} - \omega_{H} + \omega_{V}} )}\quad t}}\end{matrix} & (19) \\\begin{matrix}{{Im}_{14} = {{0.5k\quad{\sin( {\omega_{H} - \omega_{V}} )}\quad t} +}} \\{{0.5{\sin( {{2\omega_{F}} + \omega_{H} - \omega_{V}} )}\quad t} + {0.5{\sin( {{2\omega_{F}} - \omega_{H} + \omega_{V}} )}\quad t}}\end{matrix} & (20)\end{matrix}$The following equations (21) describe the quadrature-phase response Q ofthe complex multiplier 15. $\begin{matrix}\begin{matrix}{Q = {{{Re}_{14}\sin\quad\omega_{F}t} - {{Im}_{14}\cos\quad\omega_{F}t}}} \\{= {{0.5{\cos( {\omega_{H} - \omega_{V}} )}t*\sin\quad\omega_{F}t} +}} \\{{0.5{\cos( {{2\omega_{F}} + \omega_{H} - \omega_{V}} )}t*\sin\quad\omega_{F}t} +} \\{{0.5{\cos( {{2\omega_{F}} - \omega_{H} + \omega_{V}} )}t*\sin\quad\omega_{F}t} -} \\{{0.5{\sin( {\omega_{H} - \omega_{V}} )}t*\cos\quad\omega_{F}t} -} \\{{0.5{\sin( {{2\omega_{F}} + \omega_{H} - \omega_{V}} )}t*\cos\quad\omega_{F}t} -} \\{0.5{\sin( {{2\omega_{F}} - \omega_{H} + \omega_{V}} )}t*\cos\quad\omega_{F}t} \\{= {{0.5\lbrack {{{\cos( {\omega_{H} - \omega_{V}} )}t*\sin\quad\omega_{F}t} - {\sin( {\omega_{H} - \omega_{V}} )t*\cos\quad\omega_{F}t}} \rbrack} +}} \\{0.5\lbrack {{{\cos( {{2\omega_{F}} + \omega_{H} - \omega_{V}} )}t*\sin\quad\omega_{F}t} -} } \\{ {\sin( {{2\omega_{F}} + \omega_{H} - \omega_{V}} )t*\cos\quad\omega_{F}t} \rbrack +} \\{0.5\lbrack {{{\cos( {{2\omega_{F}} - \omega_{H} + \omega_{V}} )}t*\sin\quad\omega_{F}t} -} } \\ {\sin( {{2\omega_{F}} - \omega_{H} + \omega_{V}} )t*\cos\quad\omega_{F}t} \rbrack \\{= {{{+ 0.5}{\sin( {\omega_{F} - \omega_{H} + \omega_{V}} )}t} +}} \\{{0.5{\sin( {\omega_{F} + \omega_{H} - \omega_{V}} )}t} +} \\{0.5\sin( {\omega_{F} - \omega_{H} + \omega_{V}} )t} \\{= {{{\sin( {\omega_{F} - \omega_{H} + \omega_{V}} )}t} + {0.5\sin( {\omega_{F} + \omega_{H} - \omega_{V}} )t}}}\end{matrix} & (21)\end{matrix}$Presuming (ω_(H)−ω_(V)) to be approximately ω_(F), the lowpass filter 16suppresses the higher frequency cos 0.5 sin (ω_(F)+ω_(H)−ω_(V))tcomponent of the Q signal, to generate a response R₁₆ that within theAFPC bandwidth is an ensemble of terms each per the following equation(22).R ₁₆=sin (ω_(F)−ω_(H)+ω_(V))t  (22)R₁₆ is an AFPC signal that will adjust ω_(H) so that (ω_(H)−ω_(V))equals ω_(F) to reduce error signal substantially to zero.

The following equations (23) describe the in-phase response I of thecomplex multiplier 15. $\begin{matrix}\begin{matrix}{I = {{{Re}_{14}\cos\quad\omega_{F}t} - {{Im}_{14}\sin\quad\omega_{F}t}}} \\{= {{{+ 0.5}{\cos( {\omega_{H} - \omega_{V}} )}t*\cos\quad\omega_{F}t} +}} \\{{0.5{\cos( {{2\omega_{F}} + \omega_{H} - \omega_{V}} )}t*\cos\quad\omega_{F}t} +} \\{{0.5{\cos( {{2\omega_{F}} - \omega_{H} + \omega_{V}} )}t*\cos\quad\omega_{F}t} +} \\{{0.5{\sin( {\omega_{H} - \omega_{V}} )}t*\sin\quad\omega_{F}t} +} \\{{0.5{\sin( {{2\omega_{F}} + \omega_{H} - \omega_{V}} )}t*\sin\quad\omega_{F}t} +} \\{0.5{\sin( {{2\omega_{F}} - \omega_{H} + \omega_{V}} )}t*\sin\quad\omega_{F}t} \\{= {{+ {0.5\lbrack {{{\cos( {\omega_{H} - \omega_{V}} )}t*\cos\quad\omega_{F}t} + {{\sin( {\omega_{H} - \omega_{V}} )}t*\sin\quad\omega_{F}t}} \rbrack}} +}} \\{0.5\lbrack {{{\cos( {{2\omega_{F}} + \omega_{H} - \omega_{V}} )}t*\cos\quad\omega_{F}t} +} } \\{ {{\sin( {{2\omega_{F}} + \omega_{H} - \omega_{V}} )}t*\sin\quad\omega_{F}t} \rbrack +} \\{0.5\lbrack {{{\cos( {{2\omega_{F}} - \omega_{H} + \omega_{V}} )}t*\cos\quad\omega_{F}t} +} } \\ {{\sin( {{2\omega_{F}} - \omega_{H} + \omega_{V}} )}t*\sin\quad\omega_{F}t} \rbrack \\{= {{{+ 0.5}{\cos( {\omega_{F} - \omega_{H} + \omega_{V}} )}t} +}} \\{{0.5{\cos( {\omega_{F} + \omega_{H} - \omega_{V}} )}t} +} \\{0.5{\cos( {\omega_{F} - \omega_{H} + \omega_{V}} )}t} \\{= {{{\cos( {\omega_{F} - \omega_{H} + \omega_{V}} )}t} + {0.5{\cos( {\omega_{F} + \omega_{H} - \omega_{V}} )}t}}}\end{matrix} & (23)\end{matrix}$Suppose that (ω_(V)−ω_(H)) exhibits variation of higher frequency thanthe AFPC time constant. Each component of the ensemble descriptive ofthese variations is assumed to have a (ω_(H)−ω_(V)) value of(ω_(F)+ω_(M)). When the AFPC loop is phase-locked, the in-phase responseI of the complex multiplier 15 will be an ensemble of the followingcomponent I responses, as determined by substituting(ω_(F)+ω_(M))for(ω_(H)−ω_(V)) in equation (24). $\begin{matrix}\begin{matrix}{I = {{{\cos( {- \omega_{M}} )}t} + {0.5{\cos( {{2\omega_{F}} + \omega_{M}} )}\quad t}}} \\{= {{\cos\quad\omega_{M}t} + {0.5{\cos( {{2\omega_{F}} + \omega_{M}} )}\quad t}}}\end{matrix} & (24)\end{matrix}$

FIG. 7 shows in more detail how the FIG. 6 concept is implemented whenthe demodulation is carried out in the digital regime, with the phasesplitter 14 being the phase-splitter 014 of digital-filter type and thecomplex multiplier 15 being the digital complex multiplier 015. Thelowpass filter 13 response R₁₃ is digitized by the analog-to-digitalconverter 17. The resulting digitized VSB signal is applied as a firstof its two summand input signals to a digital adder 026 and is appliedas a multiplicand input signal to a digital multiplier 027. A read-onlymemory 28 is addressed in parallel with the ROM 21 and 22, which supplythe digital complex multiplier 015 sample streams respectivelydescriptive of a cos ω_(F)t system function and descriptive of a sinω_(F)t system function. The ROM 28 supplies the digital multiplier 027as the multiplier input signal thereof a sample stream descriptive of acos 2ω_(F)t system function, which is delayed to compensate for thelatent delays in the mixer 10, the analog lowpass filter 13 and the ADC17. The product output signal from the digital multiplier 027 is appliedto the digital adder 026 as the second of its two summand input signals.The sum output signal of the digital adder 026 supplied to thephase-splitter 014 as its input signal comprises a DSB AM signalcomponent.

The in-phase response I supplied by the digital complex multiplier 015as the real part of the complex product signal therefrom is a digitalbaseband signal accompanied by a sideband of the cos 2ω_(F)t carrier inaccordance with equation (24). A rate-reduction filter 29 with 2ω_(F)output sample rate receives this in-phase response I and aliases thesideband of the cos 2ω_(F)t carrier to baseband to augment the basebandsignal. The rate-reduced I response from the rate-reduction filter 29 issuited for application to subsequent portions of the receiver not shownin FIG. 7. These subsequent portions include baseband equalization andghost cancellation filtering and subsequent symbol decoder apparatus.The quadrature-phase response Q supplied by the digital complexmultiplier 015 as the imaginary part of the complex product signaltherefrom is a digital baseband signal, which a digital-to-analogconverter 19 converts to analog form to provide the input signal to theanalog lowpass filter 16 that supplies AFPC signal to the VCO 11.

FIG. 8 shows a modification of the FIG. 6 portion of a VSB radio signalreceiver that uses a complex mixer instead of the mixer 10 fordownconverting VSB VHF I-F signal to VSB final I-F signal. This avoidsthe need for the phase-splitter 14 before the complex multiplier 15 usedfor demodulation. The complex mixer comprises component mixers 100 and101 having their respective output signals filtered by lowpass filters130 and 131, respectively. The lowpass filter 130 response is applied asthe first of two summand input signals to an adder 260. The othersummand input signal of the adder 260 is the product output signal of abalanced amplitude modulator 270 which modulates a suppressed carrierfrequency 2ω_(f) in accordance with the lowpass filter 130 response. Thesum output signal that the adder 260 generates includes DSB AM of aω_(F) carrier and is supplied to the complex multiplier 15 as a realcomponent of final I-F input signal. The lowpass filter 131 response isapplied as the first of two summand input signals to an adder 261. Theother summand input signal of the adder 261 is the product output signalof a balanced amplitude modulator 271 which modulates a suppressedcarrier frequency 2ω_(f) in accordance with the lowpass filter 131response. The sum output signal that the adder 261 generates includesDSB AM of a ω_(F) carrier and is supplied to the complex multiplier 15as an imaginary component of final I-F input signal.

The mixers 100 and 101 receive similar VSB amplified VHF I-F signals asrespective multiplicand input signals to be downconverted, which VSBsignals can be supplied from the customary gain-controlled VHF I-Famplifier chain. The FIG. 6 VCO 11 supplying cos ω_(H)t real or in-phaselocal oscillations is replaced in FIG. 8 by a VCO 011 supplying sinω_(H)t imaginary or quadrature-phase local oscillations, as well assupplying cos ω_(H)t real or in-phase local oscillations. The cos ω_(H)tin-phase local oscillations from the VCO 011 are applied as multiplierinput signal to the component mixer 100. The operation of the modulator120, the mixer 100, the lowpass filter 130, the adder 260 and themultiplier 270 in the portion of a VSB signal receiver shown in FIG. 8corresponds with the operation of the modulator 12, the mixer 10, thelowpass filter 13, the adder 26 and the multiplier 27 in the portion ofa VSB signal receiver shown in FIG. 6. So, in accordance with equation(18) the sum output signal R₂₆₀ from the adder 260 is an ensemble ofterms each per the following equation (25). $\begin{matrix}\begin{matrix}{R_{260} = {R_{130} + R_{270}}} \\{{= {{0.5{\cos( {\omega_{H} - \omega_{V}} )}t} +}}\quad} \\{{0.5{\cos( {{2\omega_{F}} + \omega_{H} - \omega_{V}} )}t} + {0.5{\cos( {{2\omega_{F}} - \omega_{H} + \omega_{V}} )}t}}\end{matrix} & (25)\end{matrix}$The sin ω_(H)t in-phase local oscillations from the VCO 011 are appliedas multiplier input signal to the component mixer 101, in accordancewith the trigonometric identity (3), the response R₁₀₁ from the mixer101 will be an ensemble of terms each of the following form, presumingthe VCO 111 to be of the form cos ω_(H)t.R ₁₀₁=0.5 sin (ω_(H)−ω_(V))t+0.5 sin (ω_(H)+ω_(V))t  (26)The lowpass filter 131 suppresses the high frequency terms in itsresponse R₁₃ to the mixer 10 response R₁₀, to generate an ensemble ofterms each per the following equation (27).R ₁₃=0.5 sin (ω_(H)−ω_(V))t  (27)The balanced amplitude modulator 271 modulates a suppressed 2 cos 2ω_(F)t carrier by the lowpass filter 131 response R₁₃₁ to generate inits response R₂₇₁, in accordance with the trigonometric identity (3), anensemble of terms each per the following equation (28).R ₂₇₁=0.5 sin (2ω_(F)+ω_(H)−ω_(V))t+0.5 sin (2ω_(F)−ω_(H)+ω_(V))t  (28)The adder 261 sums R₁₃₁ and R₂₇₁ to generate a sum output signal R₂₆₁which is an ensemble of terms each per the following equation (29).$\begin{matrix}\begin{matrix}{R_{261} = {R_{131} + R_{271}}} \\{= {{0.5\quad{\sin( {\omega_{H} - \omega_{V}} )}t} +}} \\{{0.5\quad{\sin( {{2\omega_{F}} + \omega_{H} - \omega_{V}} )}t} + {0.5\quad{\sin( {{2\omega_{F}} - \omega_{H} + \omega_{V}} )}t}}\end{matrix} & (29)\end{matrix}$The adder 130 response R₁₃₀ in equation (25) and the adder 131 responseR₁₃₁ in equation (29) respectively correspond to the real response Re₁₄of the phase-splitter 14 per equation (19) and to the imaginary responseIm₁₄ of the phase-splitter 14 per equation (20).

FIG. 9 shows modifications of the FIG. 7 portion of the VSB signalreceiver to use the complex mixer composed of component mixers 100 and101 rather than the mixer 10, so that the phase-splitter 014 ofdigital-filter type is not required for supplying complex multiplicandinput signal to the digital complex multiplier 015. As in FIG. 8, FIG. 9shows the component mixers 100 and 101 having their respective outputsignals filtered by lowpass filters 130 and 131, respectively. Themixers 100 and 101 receive similar VSB amplified VHF I-F signals asrespective multiplicand input signals to be downconverted to VSB finalI-F signals, which VSB VHF I-F signals can be supplied from thecustomary gain-controlled VHF I-F amplifier chain. The FIG. 7 VCO 11supplying cos ω_(H)t real or in-phase local oscillations is replaced inFIG. 9 by the VCO 011 supplying sin ω_(H)t imaginary or quadrature-phaselocal oscillations, as well as supplying cos ω_(H)t real or in-phaselocal oscillations. The cos ω_(H)t in-phase local oscillations from theVCO 011 are supplied to the component mixer 100 as multiplier inputsignal thereto. The operation of the modulator 120, the mixer 100 andthe lowpass filter 130 in the portion of a VSB signal receiver shown inFIG. 9 corresponds with the operation of the modulator 12, the mixer 10and the lowpass filter 13 in the portion of a VSB signal receiver shownin FIG. 7. As in FIG. 8, FIG. 9 shows the sine ω_(H)t in-phase localoscillations from the VCO 011 being supplied to the component mixer 101as multiplier input signal thereto.

The responses R₁₃₀ and R₁₃₁ of the lowpass filters 130 and 131 aredigitized by analog-to-digital converters 170 and 171, respectively. Thedigitized VSB signal from the ADC 170 is applied as a first of its twosummand input signals to a digital adder 0260 and is applied as amultiplicand input signal to a digital multiplier 0270. The digitizedVSB signal from the ADC 171 is applied as a first of its two summandinput signals to a digital adder 0261 and is applied as a multiplicandinput signal to a digital multiplier 0271. The ROMs 21, 22 and 28 in theFIG. 9 apparatus are connected and operated the same as in the FIG. 7apparatus. The ROM 28 supplies a sample stream descriptive of a 2 cos2ω_(F)t system function applied to the digital multipliers 0270 and 0271as the multiplier input signals thereof. The sum output signals from thedigital adders 0260 and 0261 are supplied to the digital complexmultiplier 015 as real and imaginary signals, respectively.

The in-phase response I supplied by the digital complex multiplier 015as the real part of the complex product signal therefrom is a digitalbaseband signal accompanied by a sideband of the cos 2ω_(F)t carrier inaccordance with equation (24). A rate-reduction filter 29 with 2ω_(F)output sample rate receives this in-phase response I and aliases thesideband of the cos 2ω_(F)t carrier to baseband to augment the basebandsignal. The rate-reduced I response from the rate-reduction filter 29 issuited for application to subsequent portions of the receiver not shownin FIG. 7. These subsequent portions include baseband equalization andghost cancellation filtering and subsequent symbol decoder apparatus.The quadrature-phase response Q supplied by the digital complexmultiplier 015 as the imaginary part of the complex product signaltherefrom is a digital baseband signal, which the digital-to-analogconverter 19 converts to analog form to provide the input signal to theanalog lowpass filter 16 that supplies AFPC signal to the VCO 011.

FIG. 10 shows the FIG. 7 apparatus for demodulating a VSB DTV signal, asmodified to include NTSC sound trap filtering 30 of the final I-F signalbefore the ADC 17. FIG. 10 also shows a buffer amplifier 31 for applyingthe response of the analog lowpass filter 13 to the NTSC sound trapfiltering 30. The buffer amplifier 31 presents high input impedanceloading on the analog lowpass filter 13 and low output impedance assource impedance for the NTSC sound trap filtering 30. The NTSC soundtrap filtering 30 can comprise a trap filter for co-channel interferingNTSC sound carrier, or a trap filter for adjacent channel NTSC soundcarrier, or trap filters for both types of NTSC sound carrier. Since theNTSC sound carriers accompanying the final I-F signal are more than 5megahertz above zero frequency when the mixer 101 is designed to recoveran upper sideband of the VSB signal as downconverted to final I-F band,the trap filters can be configurations such as bridged-tee filters thatare known in analog television receiver design.

FIG. 11 shows NTSC trap filtering 300 as described by A. L. R. Limbergand C. B. Patel in their U.S. Pat. No. 6,496,230 titled “DIGITAL TVSIGNAL RECEIVER WITH DIRECT CONVERSION FROM UHF I-F TO LOW-BAND I-FBEFORE DIGITAL DEMODULATION”. The FIG. 10 apparatus can be modified toreplace the NTSC sound trap filtering 30 with the NTSC trap filtering300. The NTSC trap filtering 300 suppress portions of any co-channelinterfering NTSC signal proximate the video carrier, as well as portionsof any co-channel interfering NTSC signal proximate the audio carrier.The NTSC trap filtering 300, which comprises elements 301-310, ispreferred for the ease with which it can be disabled when there is nosignificant level of co-channel interfering NTSC signal. The low-bandI-F buffer amplifier 31 output signal is applied to the non-invertinginput terminal of a differential-input amplifier 310, the low-band I-Foutput signal of which amplifier 310 is digitized by the ADC 17 of FIG.10.

FIG. 11 shows the low-band buffer amplifier 31 connected for supplyingits output signal to a ceramic bandpass filter 301, which selectivelyresponds to a frequency range including the audio carrier of theco-channel interfering NTSC signal, and to a ceramic bandpass filter302, which selectively responds to a frequency range including the videocarrier of the co-channel interfering NTSC signal. The frequency rangeto which the ceramic bandpass filter 301 selectively responds can, forexample, extend about ±50 kilohertz each side of the audio carrier ofthe co-channel interfering NTSC signal as translated to the low-bandfinal I-F band. By way of further example, the frequency range that theceramic bandpass filter 301 selectively responds to can be extendedsomewhat further from the NTSC audio carrier towards the edge of thereception channel closest by the NTSC audio carrier.

The frequency range to which the ceramic bandpass filter 302 selectivelyresponds should be within a frequency range substantially within a widerfrequency range extending 896.9 to 978.4 kHz from DTV carrier astranslated to the low-band final I-F band. This avoids the ceramicbandpass filter 302 response including any subharmonic of symbol rate,since the tenth and eleventh subharmonics respectively fall 978.4 kHzand 896.9 kHz from DTV carrier.

The ceramic bandpass filter 301 response is applied as input signal to avoltage amplifier 303, and the ceramic bandpass filter 302 response isapplied as input signal to a voltage amplifier 304. An analog adder 305sums the responses of the voltage amplifiers 303 and 304 to generate asum signal that a transmission gate 306 selectively applies to theinverting input terminal of the differential-input amplifier 310. Thevoltage gain of the voltage amplifier 303 is chosen to compensate forinsertion losses for the signal passed through the ceramic bandpassfilter 301, the adder 305, and the conductive transmission gate 306. Thevoltage gain of the voltage amplifier 30 is chosen to compensate forinsertion losses for the signal passed through the ceramic bandpassfilter 302, the adder 305, and the conductive transmission gate 306.

The transmission gate 306 is rendered conductive by a co-channel NTSCinterference detector 307 supplying an indication that there is aco-channel interfering NTSC signal of enough energy to significantlyaffect data slicing and other symbol decoding procedures. The co-channelNTSC interference detector 307 can take a number of forms, but apreferred form multiplicatively mixes the responses of the ceramicbandpass filters 301 and 302 one with the other, which generates acontinuous 4.5 MHz intercarrier signal whenever co-channel interferingNTSC signal is present in the low-band I-F buffer amplifier 31 outputsignal. As a practical consideration, the 4.5 MHz intercarrier signal isnot generated when only DTV signal is being received. A bandpass filterselects the 4.5 MHz intercarrier signal for envelope detection, and theenvelope detection result is threshold detected for determining whetheror not a 4.5 MHz intercarrier signal of significant energy results frommultiplicatively mixing the responses of the ceramic bandpass filters301 and 302 one with the other.

The co-channel NTSC interference detector 307 indications are suppliedto a logic inverter 308, the response of which controls transmissionthrough a transmission gate 309. The transmission gate 309 is renderednon-conductive when the co-channel NTSC interference detector 307supplies an indication that there is a co-channel interfering NTSCsignal of enough energy to significantly affect data slicing and othersymbol decoding procedures. The concurrent conduction of thetransmission gate accordingly 306 applies to the inverting inputterminal of the differential-input amplifier 310 a signal correspondingto the portions of the low-band I-F buffer amplifier 31 output signal inthe frequency regions near the NTSC audio carrier and near the NTSCvideo carrier. Shimming delay is included in the FIG. 11 circuitry sothat the low-band I-F buffer amplifier 31 output signal applied to thenon-inverting input terminal of the differential-input amplifier 310 isdelayed similarly to the responses of the responses of the ceramicbandpass filters 301 and 302 as selectively applied to the invertinginput terminal of the differential-input amplifier 310. Accordingly, thedifferential-input amplifier 310 exhibits suppressed response to theportions of the buffer amplifier 31 output signal in the frequencyregions near the NTSC audio carrier and near the NTSC video carrier, ascompared to the response to other portions of the low-band I-F bufferamplifier 31 output signal.

The transmission gate 306 is rendered non-conductive by the co-channelNTSC interference detector 307 supplying an indication that there is noco-channel interfering NTSC signal with enough energy to significantlyaffect data slicing and other symbol decoding procedures. Thisindication renders the transmission gate 309 conductive to apply areference direct potential to the inverting input terminal of thedifferential-input amplifier 310. Accordingly, the differential-inputamplifier 310 exhibits response to the entire low-band I-F bufferamplifier 31 output signal. That is, if there is no co-channelinterfering NTSC signal with enough energy to significantly affect dataslicing and other symbol decoding procedures, the DTV signal is notsubjected to trap filtering.

Thusfar, it has been presumed that the shaping of the channel responseof the receiver in the carrier-frequency region is accomplishedprimarily in the UHF or VHF intermediate-frequency amplifiers precedingthe mixer 10 used for downconverting to the final I-F band. Insofar asin-phase demodulation of the DSB AM DTV signal is concerned, it isdesirable that the DTV receiver introduce roll-off through thecarrier-frequency region to augment by an additional 3 dB the 3 dBroll-off introduced at the DTV transmitter. This results in an overallchannel response which after demodulation is nominally flat down to zerofrequency, reducing the amount of equalization that must be introducedat these frequencies. However, insofar as quadrature-phase demodulationof the DSB AM DTV signal is concerned, it is preferable not to roll offthe I-F amplifier responses in the carrier frequency region. Phaseresponse is less affected in the carrier-frequency region if furtherroll-off of channel response in this region is avoided, although theVSB-to-DSB-AM conversion techniques of the invention substantially avoidthis deleterious effect. Avoiding farther roll-off of channel responsein the carrier-frequency region avoids some loss of carrier-to-noiseratio caused by quantization noise introduced during digitization of thefinal I-F signal.

FIG. 12 shows modification to the FIG. 3 circuitry in regard toprocessing the digitized final I-F signal from the analog-to-digitalconverter 17, which modification is useful if further roll-off ofchannel response in the carrier-frequency region is avoided in theintermediate-frequency amplifier chain. The digital complex multiplier015 is replaced by a modified digital complex multiplier 150, andphase-splitter filtering 141 augments the phase-splitter 014. Themodified digital complex multiplier 150 receives real and imaginarystreams of samples from the phase-splitter 014 as input signal for theportion of that modified complex multiplier comprising elements 151-153that generates the quadrature-phase (Q) portion of the complex productinput signal, but not for the portion of that modified complexmultiplier comprising elements 154-156 that generates the in-phase (I)portion of the complex product. Instead, the modified digital complexmultiplier 150 receives real and imaginary streams of samples fromphase-splitting filtering 141 as input signal for the portion of thatmodified complex multiplier comprising elements 154-156 that generatesthe in-phase (I) portion of the complex product. The phase-splitterfiltering 141 exhibits a dip in system function at the middle of thefinal I-F band to provide an amplitude-versus-frequency response thatpreferably is flat through the region of carrier frequency insofar asthe overall channel response is concerned.

Digital multipliers 151, 152, 154 and 155 are included within themodified digital complex multiplier 150. In order that the latent delayin generating product signals be minimized, the digital multipliers 151,152, 154 and 155 are preferably constructed using read-only memory,rather than using logic circuitry and registers for multiplier andmultiplicand signals. The digital multiplier 151 multiplies theimaginary component of the digitized final I-F signal from thephase-splitter 014 by the real component of the complex digital carrierread from the ROM 20. The digital multiplier 152 multiplies the realcomponent of the digitized final I-F signal from the phase-splitter 014by the imaginary component of the complex digital carrier read from theROM 22. The digital adder 153 sums the product output signals from thedigital multipliers 151 and 52 to generate a sum output signal suppliedas the quadrature-phase (Q) baseband output signal from the modifieddigital complex multiplier 150. The digital multiplier 154 multipliesthe real component of the digitized final I-F signal from thephase-splitter filtering circuitry 141 by the real component of thecomplex digital carrier read from the ROM 20. The digital multiplier 155multiplies the imaginary component of the digitized final I-F signalfrom the phase-splitter filtering circuitry 141 by the imaginarycomponent of the complex digital carrier read from the ROM 22. Thedigital subtractor 156 differentially combines the product outputsignals from the digital multipliers 154 and 155 to generate adifference output signal supplied as the in-phase (I) baseband outputsignal from the modified digital complex multiplier 150.

Further modifications of the modified digital complex multiplier 150reduce the amount of ROM required overall, but provide equivalentfunction insofar as synchrodyning DSB AM signal to baseband isconcerned. In these further modifications the digital multipliers 151,152, 154 and 155 are replaced by ROMs directly addressed from theaddress generator previously used for addressing the ROMs 20 and 22, andthe ROMs 20 and 22 are dispensed with. The digital complex multiplier015 can also be modified to use this reduced-ROM structure.

While the invention has been described in the particular context of DTVreceivers, it should be appreciated that the invention is useful, aswell, for the reception of VSB radio signals used in other types ofcommunications.

1. An apparatus for practicing a method for extracting information froma vestigial sideband amplitude-modulation signal generated in accordancewith a modulating signal, said method for extracting information fromsaid vestigial sideband amplitude-modulation signal comprising the stepsof: mixing said vestigial sideband amplitude-modulation signal with afirst beat frequency different from a earner frequency of said vestigialsideband amplitude-modulation signal, said first beat frequency being ofsuch value as to generate a first mixing result that is translated infrequency to have a carrier at an offset frequency; mixing saidvestigial sideband amplitude-modulation signal with a second beatfrequency different from the carrier frequency of said vestigialsideband amplitude-modulation signal and from said first beat frequency,said second beat frequency being of such value as to generate a secondmixing result that is translated in frequency to have a carrier at saidoffset frequency; and combining said first and second mixing results,which have respective frequency spectra that are images of each other,to form said double-sideband amplitude-modulation signal with itscarrier frequency at said offset frequency; for converting saidvestigial sideband amplitude-modulation signal to said double-sidebandamplitude-modulation signal containing similar information, and saidmethod for extracting information from said vestigial sidebandamplitude-modulation signal further comprising the step of detecting theinformation contained in said double-sideband amplitude-modulationsignal, said apparatus comprising: a local oscillator for generatinglocal oscillations of a frequency and phase controlled by anautomatic-frequency-and-phase-control signal; a balanced modulatorconnected for amplitude-modulating said local oscillations in accordancewith said offset frequency, to generate a balanced modulator outputsignal essentially consisting of a first beat frequency component and asecond beat frequency component; a mixer connected for mixing saidbalanced modulator output signal with said vestigial sidebandamplitude-modulation signal, to generate a mixer output signal includingsaid double-sideband amplitude-modulation signal; an image-rejectionfilter connected for supplying said double-sideband amplitude-modulationsignal as a selective response to said mixer output signal; aphase-splitter connected for generating a complex response to saiddouble-sideband amplitude-modulation signal said image-rejection filtersupplies as a selective response to said mixer output signal; a complexmultiplier connected for multiplying said complex response to saiddouble-sideband amplitude-modulation signal by a complex carrier thefrequency of which equals said offset frequency, to generate an in-phasesynchronous detection response and a quadrature-phase synchronousdetection response; and circuitry for generating saidautomatic-frequency-and-phase-control signal from said quadrature-phasesynchronous detection response supplied thereto, whichautomatic-frequency-and-phase-control signal is applied to said localoscillator to complete an automatic-frequency-and-phase-control loop foradjusting the frequency and phase of the local oscillations generated bysaid local oscillator so that said in-phase synchronous detectionresponse reproduces the modulating signal in accordance with which saidvestigial sideband amplitude-modulation signal was generated.
 2. Theapparatus of claim 1, wherein said circuitry for generating saidautomatic-frequency-and-phase-control signal from said quadrature-phasesynchronous detection response supplied thereto comprises an analoglowpass filter connected for supplying said local oscillator saidautomatic-frequency-and-phase-control signal in response to saidquadrature-phase synchronous detection response.
 3. An apparatus forpracticing a method for extracting information from a vestigial sidebandamplitude-modulation signal generated in accordance with a modulatingsignal, said method for extracting information from said vestigialsideband amplitude-modulation signal comprising the steps of: mixingsaid vestigial sideband amplitude-modulation signal with a first beatfrequency different from a carrier frequency of said vestigial sidebandamplitude-modulation signal, said first beat frequency being of suchvalue as to generate a first mixing result that is translated infrequency to have a carrier at an offset frequency; mixing saidvestigial sideband amplitude-modulation signal with a second beatfrequency different from the carrier frequency of said vestigialsideband amplitude-modulation signal and from said first beat frequency,said second beat frequency being of such value as to generate a secondmixing result that is translated in frequency to have a carrier at saidoffset frequency; and combining said first and second mixing results,which have respective frequency spectra that are images of each other,to form said double-sideband amplitude-modulation signal with itscarrier frequency at said offset frequency; for converting saidvestigial sideband amplitude-modulation signal to said double-sidebandamplitude-modulation signal containing similar information, and saidmethod for extracting information from said vestigial sidebandamplitude-modulation signal further comprising the step of detecting theinformation contained in said double-sideband amplitude-modulationsignal, said apparatus comprising: a local oscillator for generatinglocal oscillations of a frequency and phase controlled by anautomatic-frequency-and-phase-control signal; a balanced modulatorconnected for amplitude-modulating said first local oscillations inaccordance with said offset frequency, to generate a balanced modulatoroutput signal essentially consisting of a first beat frequency componentand a second beat frequency component; a mixer connected for mixing saidbalanced modulator output signal with said vestigial sidebandamplitude-modulation signal, to generate a mixer output signal includingsaid double-sideband amplitude-modulation signal; an image-rejectionfilter connected for supplying said double-sideband amplitude-modulationsignal as a selective response to said mixer output signal; ananalog-to-digital converter for digitizing the selective response ofsaid image-rejection filter to supply a digitized double-sidebandamplitude-modulation signal; a digital phase-splitter connected forgenerating a complex response to said digitized double-sidebandamplitude-modulation signal; a digital complex multiplier connected formultiplying said complex response to said digitized double-sidebandamplitude-modulation signal by a complex digital carrier the frequencyof which equals said offset frequency, to generate a digital in-phasesynchronous detection response and a digital quadrature-phasesynchronous detection response; and circuitry for generating saidautomatic-frequency-and-phase-control signal from said digitalquadrature-phase synchronous detection response supplied thereto, whichautomatic-frequency-and-phase-control signal is applied to said localoscillator to complete an automatic-frequency-and-phase-control loop foradjusting the frequency and phase of the local oscillations generated bysaid local oscillator so that said digital in-phase synchronousdetection response describes the modulating signal in accordance withwhich said vestigial sideband amplitude-modulation signal was generated.4. The apparatus of claim 3, wherein said circuitry for generating saidautomatic-frequency-and-phase-control signal from said digitalquadrature-phase synchronous detection response supplied theretocomprises: a digital-to-analog converter connected for responding tosaid digital quadrature-phase synchronous detection response to supply adigital-to-analog converter response; and an analog lowpass filterconnected for supplying said local oscillator saidautomatic-frequency-and-phase-control signal in response to saiddigital-to-analog converter response.
 5. The apparatus of claim 3,further comprising: read-only memory, for supplying said digital complexmultiplier with said complex digital carrier to multiply said complexresponse to said digitized double-sideband amplitude-modulation signalby, and for supplying a digitized offset frequency signal.
 6. Anapparatus for practicing a method for extracting information from avestigial sideband amplitude-modulation signal generated in accordancewith a modulating signal, said method for extracting information fromsaid vestigial sideband amplitude-modulation signal comprising the stepsof: mixing said vestigial sideband amplitude-modulation signal with afirst beat frequency different from a carrier frequency of saidvestigial sideband amplitude-modulation signal, said first beatfrequency being of such value as to generate a first mixing result thatis translated in frequency to have a carrier at an offset frequency;mixing said vestigial sideband amplitude-modulation signal with a secondbeat frequency different from the carrier frequency of said vestigialsideband amplitude-modulation signal and from said first beat frequency,said second beat frequency being of such value as to generate a secondmixing result that is translated in frequency to have a carrier at saidoffset frequency; and combining said first and second mixing results,which have respective frequency spectra that are images of each other,to form said double-sideband amplitude-modulation signal with itscarrier frequency at said offset frequency; for converting saidvestigial sideband amplitude-modulation signal to said double-sidebandamplitude-modulation signal containing similar information, and saidmethod for extracting information from said vestigial sidebandamplitude-modulation signal further comprising the step of detecting theinformation contained in said double-sideband amplitude-modulationsignal, said apparatus comprising: a complex multiplier for multiplyinga complex double-sideband amplitude-modulation signal by a complexcarrier the frequency of which equals said offset frequency, to generatean in-phase synchronous detection response and a quadrature-phasesynchronous detection response; a local oscillator for generatingorthogonal first and second phases of local oscillations, the frequencyand phases of which are controlled by anautomatic-frequency-and-phase-control signal; a first balanced modulatorconnected for amplitude-modulating said first phase of localoscillations in accordance with said offset frequency, to generate afirst balanced modulator output signal essentially consisting of a firstbeat frequency component and a second beat frequency component; a firstmixer connected for mixing said first balanced modulator output signalwith said vestigial sideband amplitude-modulation signal, to generate afirst mixer output signal including a real component of said complexdouble-sideband amplitude-modulation signal; a first image-rejectionfilter connected for supplying to said digital complex multiplier saidreal component of said complex double-sideband amplitude-modulationsignal as a selective response to said first mixer output signal; asecond balanced modulator connected for amplitude-modulating said secondphase of local oscillations in accordance with said offset frequency, togenerate a second balanced modulator output signal essentiallyconsisting of a third beat frequency component and a fourth beatfrequency component, said first and third beat frequency componentsbeing the same in frequency but orthogonal in phase, and said second andfourth beat frequency components being the same in frequency butorthogonal in phase; a second mixer connected for mixing said secondbalanced modulator output signal with said vestigial sidebandamplitude-modulation signal, to generate a second mixer output signalincluding an imaginary component of said complex double-sidebandamplitude-modulation signal; a second image-rejection filter connectedfor supplying to said digital complex multiplier said imaginarycomponent of said complex double-sideband amplitude-modulation signal asa selective response to said second mixer output signal; and circuitryfor generating said automatic-frequency-and-phase-control signal fromsaid quadrature-phase synchronous detection response, which circuitrycompletes an automatic-frequency-and-phase-control loop adjusting thefrequency and phase of the local oscillations generated by said localoscillator so that said in-phase synchronous detection responsereproduces the modulating signal in accordance with which said vestigialsideband amplitude-modulation signal was generated.
 7. The apparatus ofclaim 6, wherein said circuitry for generating saidautomatic-frequency-and-phase-control signal from said quadrature-phasesynchronous detection response supplied thereto comprises an analoglowpass filter connected for supplying said local oscillator saidautomatic-frequency-and-phase-control signal in response to saidquadrature-phase synchronous detection response.
 8. An apparatus forpracticing a method for extracting information from a vestigial sidebandamplitude-modulation signal generated in accordance with a modulatingsignal, said method for extracting information from said vestigialsideband amplitude-modulation signal comprising the steps of: mixingsaid vestigial sideband amplitude-modulation signal with a first beatfrequency different from a carrier frequency of said vestigial sidebandamplitude-modulation signal, said first beat frequency being of suchvalue as to generate a first mixing result that is translated infrequency to have a carrier at an offset frequency; mixing saidvestigial sideband amplitude-modulation signal with a second beatfrequency different from the carrier frequency of said vestigialsideband amplitude-modulation signal and from said first beat frequency,said second beat frequency being of such value as to generate a secondmixing result that is translated in frequency to have a carrier at saidoffset frequency; and combining said first and second mixing results,which have respective frequency spectra that are images of each other,to form said double-sideband amplitude-modulation signal with itscarrier frequency at said offset frequency; for converting saidvestigial sideband amplitude-modulation signal to said double-sidebandamplitude-modulation signal containing similar information, and saidmethod for extracting information from said vestigial sidebandamplitude-modulation signal further comprising the step of detecting theinformation contained in said double-sideband amplitude-modulationsignal, said apparatus comprising: a digital complex multiplierconnected for multiplying a complex digitized double-sidebandamplitude-modulation signal by a complex digital carrier, the frequencyof which equals said offset frequency, to generate a digital in-phasesynchronous detection response and a digital quadrature-phasesynchronous detection response; a local oscillator for generatingorthogonal first and second phases of local oscillations, the frequencyand phases of which are controlled by anautomatic-frequency-and-phase-control signal; a first balanced modulatorconnected for amplitude-modulating said first phase of localoscillations in accordance with said offset frequency, to generate afirst balanced modulator output signal essentially consisting of a firstbeat frequency component and a second beat frequency component; a firstmixer connected for mixing said first balanced modulator output signalwith said vestigial sideband amplitude-modulation signal, to generate afirst mixer output signal; a first image-rejection filter connected forsupplying a real double-sideband amplitude-modulation signal as aselective response to said first mixer output signal; a firstanalog-to-digital converter connected for digitizing the realdouble-sideband amplitude-modulation signal said first image-rejectionfilter supplies as a selective response to said first mixer outputsignal, said first analog-to-digital converter connected for supplyingthe digitized real double-sideband amplitude-modulation signal to saiddigital complex multiplier as a component of said complex digitizeddouble-sideband amplitude-modulation signal said digital complexmultiplier multiplies by said complex digital carrier; a second balancedmodulator connected for amplitude-modulating said second phase of localoscillations in accordance with said offset frequency, to generate asecond balanced modulator output signal essentially consisting of athird beat frequency component and a fourth beat frequency component,said first and third beat frequency components being the same infrequency but orthogonal in phase, and said second and fourth beatfrequency components being the same in frequency but orthogonal inphase; a second mixer connected for mixing said second balancedmodulator output signal with said vestigial sidebandamplitude-modulation signal, to generate a second mixer output signal; asecond image-rejection filter connected for supplying an imaginarydouble-sideband amplitude-modulation signal as a selective response tosaid second mixer output signal; a second analog-to-digital converterconnected for digitizing the imaginary double-sidebandamplitude-modulation signal said second image-rejection filter suppliesas a selective response to said second mixer output signal, said secondanalog-to-digital converter connected for supplying the digitizedimaginary double-sideband amplitude-modulation signal to said digitalcomplex multiplier as a component of said complex digitizeddouble-sideband amplitude-modulation signal said digital complexmultiplier multiplies by said complex digital carrier; and circuitry forgenerating said automatic-frequency-and-phase-control signal from saidquadrature-phase synchronous detection response, which circuitrycompletes an automatic-frequency-and-phase-control loop adjusting thefrequency and phase of the local oscillations generated by said localoscillator so that said in-phase synchronous detection responsereproduces the modulating signal in accordance with which said vestigialsideband amplitude-modulation signal was generated.
 9. The apparatus ofclaim 8, wherein said circuitry for generating saidautomatic-frequency-and-phase-control signal from said digitalquadrature-phase synchronous detection response supplied theretocomprises: a digital-to-analog converter connected for responding tosaid digital quadrature-phase synchronous detection response to supply adigital-to-analog converter response; and an analog lowpass filterconnected for supplying said local oscillator saidautomatic-frequency-and-phase-control signal in response to saiddigital-to-analog converter response.